(1) Field of the Invention
This invention relates to a strain sensor using semiconductor strain gauges for converting a mechanical strain into an electric signal as well as a drive circuit and an amplifier circuit for the same strain sensor, and more particularly to a strain sensor, which includes compensation means for compensating for the temperature dependency of the output voltage in the absence of signal and also compensation means for compensating for the temperature dependency of the detection sensitivity, as well as a drive circuit and an amplifier circuit for the same strain sensor.
(2) Description of the Related Art
Strain sensors are finding various industrial applications for converting various physical quantities such as forces, displacements, vibrations, shocks, pressures, etc. into electric quantities. Particularly, strain sensors using semiconductor strain gauges are finding rapidly increasing applications in pressure measurement, acceleration measurement, etc. because they are capable of making mass production, price reduction and size reduction since the circuit formation, shape processing and batch processing involved can be carried out all by semiconductor processing. Further, since it is possible today to carry out signal analysis inexpensively by using microcomputers, the applications have expanded to such fields as various control processes, in which a microcomputer reads out the strain sensor output signal as a digital signal through an analog-to-digital (AD) converter and executes analysis of the signal.
A strain sensor, in which semiconductor strain gauges utilizing the piezoelectric resistance effect of semiconductor are used as strain sensor elements, has high detection sensitivity and enables to make size reduction and mass production by using semiconductor processing. On the demerit side, however, the detection sensitivity varies greatly with temperature. In addition, the output voltage in the absence of signal is subject to great variations with temperature. Hitherto, there have been proposed various methods of compensation for such characteristic variations with temperature.
A known method of converting strain into electric quantity with a Wheatstone bridge having semiconductor strain gauges, will now be described with reference to FIG. 1. Referring to FIG. 1, there is shown a Wheatstone bridge 5 having semiconductor strain gauges 1 to 4. In this Wheatstone bridge circuit, the semiconductor strain gauges 1 to 4 are arranged such that with a strain given to each of them the resistances of the strain gauges 2 and 3 are changed inversely to the changes in the resistances of the strain gauges 1 and 4. The Wheatstone bridge 5 is supplied with a voltage E from a power supply source 7. When the resistances of the individual sides lose balance due to an impressed strain, a detection voltage corresponding to the resistance change of each side appears between detection voltage terminals 6. Denoting the current supplied from the power supply source 7 to the Wheatstone bridge 5 by I, the resistance of each semiconductor strain gauge in the strain-free state by R, the resistance change due to impressed stress by AR, the detection voltage S that appears between the detection voltage terminals 6 of the Wheatstone bridge 5 in correspondence to the impressed strain is EQU S=I.times..DELTA.R EQU or EQU S=E.times.(.DELTA.R/R) (1)
The resistance R of the semiconductor strain gauge is given in detail as follows: EQU R=R.sub.0 (1+.alpha.T){1+.sigma..multidot. (1+.beta.)T} (2)
where R.sub.0 represents the resistance in the strain-free state at a predetermined temperature, .alpha. represents the temperature coefficient of the resistance, .sigma. represents the stress generated in the strain gauge by the strain, represents the piezoelectric resistance coefficient, .beta. represents the temperature coefficient of the piezoelectric resistance coefficient, and T represents the ambient temperature. The piezoelectric resistance coefficient is stringently a tensor quantity and varies in dependence on the angle between the crystal orientation of the semiconductor crystal and the stress. Usually, however, the strain gauges are arranged such that the stress to be measured acts in the direction in which the piezoelectric resistance coefficient is maximum. By developing the equation (2) and ignoring the second temperature term, the resistance R will be EQU R=R.sub.0 (1+.alpha.T)+R.sub.0 {1+(.alpha.+.beta.)T}.sigma..multidot. (3)
The first term of the right side of the equation (3) represents the resistance change of the strain gauge with temperature, and the second term represents the resistance change of the strain gauge due to strain. Both .alpha. and .beta. vary with the impurity concentration in the crystal of the semiconductor strain gauges. In the case of crystalline silicon, .alpha. is in the order of several hundreds to 3,000 ppm/.degree.C., and .beta. is in the order of 1,000 to 3,000 ppm/.degree.C. In the strain-to-electricity conversion using semiconductor strain gauges, therefore, the resistance of the strain gauge has a temperature characteristic as given by the equation (3). This dictates temperature characteristic compensation in the following two aspects.
1) There are variations in the gauge resistance changes with temperature as represented by the first term of the right side of the equation (3) among the gauge resistances of the individual bridge sides, so that the output voltage varies with temperature in the absence of signal. PA1 2) The gauge resistance change due to strain as represented by the second term of the right side of the equation (3) varies with temperature, that is, the detection sensitivity varies with temperature. PA1 a) When the power supply voltage supplied to the sensor is varied, the rate of change in the detection sensitivity becomes higher than the rate of change in the power supply voltage. PA1 b) Because of the AC coupling of the amplifier circuits, a time as determined by the time constant of charging of the coupling capacitor is necessary until the operation is stabilized after the power source has been connected.
The signal-free output voltage variations with temperature variations in 1) above are attributable to the fact that the temperature coefficients and polarity are respectively different among individual Wheatstone bridges. With respect to these variations, individual sensors require adjustment that is matched to their characteristics irrespective of any compensation system that may be adopted. This is a significant cause of hindrance of industrialization. Accordingly, in applications where the DC signal detection is unnecessary, a method of making AC coupling of the amplifier circuit is generally adopted in order to prevent propagation of the Wheatstone bridge output variations in the absence of signal to the sensor output because this method is most readily adoptable and inexpensive. In this case, only the temperature characteristic of the output in the absence of signal that is based on the amplifier circuit characteristics, has effects on the sensor output in the absence of signal, and a practically sufficiently small variation characteristic is very readily obtainable.
As for the detection sensitivity variations with temperature in 2) above, the Wheatstone bridge drive voltage is changed with temperature.
FIG. 2 is a circuit diagram showing a prior art example of strain sensor, in which compensation for the detection sensitivity variations with temperature and compensation for the output voltage variations in the absence of signal with temperature as noted above are provided. Referring to FIG. 2, a Wheatstone bridge drive circuit 81 having a transistor Q64 and resistors R69 and R70 has a function of varying a voltage V.sub.81 applied to a Wheatstone bridge 5 with temperature. Denoting the power supply voltage by V.sub.CC, the base-emitter voltage of the transistor Q64 by V.sub.BE64 and the resistances of the resistors R69 and R70 by R.sub.69 and R.sub.70, the drive voltage V.sub.81 for the Wheatstone bridge 5 is given as follows: EQU V.sub.81 =V.sub.CC -{(R.sub.69 +R.sub.70)/R.sub.70 }.times.V.sub.BE64( 4)
This drive voltage V81 has a temperature characteristic with a positive temperature coefficient, such as an example of temperature characteristic as shown in FIG. 3, in which R.sub.69 =10 k.OMEGA., R.sub.70 =5 k.OMEGA., V.sub.CC =5 V. The temperature coefficient of the piezoelectric resistance coefficient is negative, and the temperature coefficient of the voltage V.sub.81 is adjustable by varying the ratio between the resistances R.sub.69 and R.sub.70. Thus, it is possible to compensate for the detection sensitivity variations with temperature of the Wheatstone bridge 5 by adjusting the temperature coefficient of the Wheatstone bridge drive voltage V.sub.81.
Now, the amplifier circuit section shown in FIG. 2 will be described. The output signal of the Wheatstone bridge 5 is amplified in a first amplifier stage 60 which has operational amplifiers 65 and 66 and resistors R72 to R74, and is further amplified in a second amplifier stage 61 which has an operational amplifier 67 and resistors R75 to R77. Then, its components at frequencies lower than a predetermined frequency are blocked by a high-pass filter 62, which has a capacitor C80 and resistors R78 and R79, and then it is coupled through an operational amplifier 68 to an output terminal 8. A variable resistor R71 is provided for adjusting the output voltage of the Wheatstone bridge 5 in the absence of signal at a predetermined initial ambient temperature to 0 V.
The first amplifier stage 60 is generally called an instrumentation amplifier with the amplification factor G.sub.60 given as EQU G.sub.60 =(R.sub.72 +R.sub.73 +R.sub.74)/R.sub.72 ( 5)
where R.sub.72 to R.sub.74 represent resistances of the resistors R72, R73 and R74. The second amplifier stage 61 serves as an inverting amplifier with its amplification factor G.sub.61 given as EQU G.sub.61 =R.sub.77 /R.sub.75 ( 6)
where R.sub.75 and R.sub.77 represent resistances of the resistors R75 and R77. The low cut-off frequency f.sub.CL62 of the high-pass filter 62 is given as EQU F.sub.CL62 =(R.sub.78 +R.sub.79)/(C.sub.80 .times.R.sub.78 .times.R.sub.79)(7)
where C.sub.80 represents the electrostatic capacitance of the capacitor C80 and R.sub.78 and R.sub.79 represent the resistances of the resistors R78 and R79. Thus, by appropriately selecting the respective values of C.sub.80, R.sub.78 and R.sub.79, it is possible to obtain great attenuation of signal components at frequencies lower than a desired frequency. Thus, very low frequency signal components such as the output voltage variations of the Wheatstone bridge 5 in the absence of signal due to temperature variations, time-involved variations, etc. can be blocked so that they do not appear at all at the output terminal 8.
As shown above, the strain sensor shown in FIG. 2 comprises standard operational amplifiers, a transistor and passive elements, and this can be readily realized. In addition, the detection sensitivity variations due to the temperature dependency characteristic of the semiconductor strain gauges can be satisfactorily compensated for. Further, since the amplifier circuits are AC coupled therebetween, there is no need at all of taking the Wheatstone bridge output voltage variations in the absence of signal into considerations. Such a circuit construction, although it cannot be used for the measurement of pressure, mass, etc. that requires measurements of DC signal components, has a merit in that the circuit is simple for such applications as vibration measurement or the like where there is no need of DC signal detection.
However, the above prior art strain sensor has the following drawbacks.
For example, with the strain sensor shown in FIG. 2, when the power supply voltage V.sub.CC is increased by 5%, the rate of change .DELTA.V.sub.81 in the drive voltage V.sub.81 for the Wheatstone bridge 5 is expressed as follows: ##EQU1## Thus, assuming here, for instance, EQU V.sub.CC =5 V, {(R.sub.69 +R.sub.70)/R.sub.70 }.times.V.sub.BE64 =1.4 V,
.DELTA.V.sub.81 is about 1.07. Thus, when the detection sensitivity is normalized with the power supply voltage, an increase of the power supply voltage V.sub.CC by 5% corresponds to an increase of the normalized sensitivity by about 2%.
The voltage V.sub.68 at the output terminal 8 in FIG. 2 in the absence of signal is expressed as follows: EQU V.sub.68 ={R.sub.79 /(R.sub.78 +R.sub.79)}.times.V.sub.CC ( 9)
Immediately upon the sensor power source connection, that is, switched-on, the mid point voltage of the Wheatstone bridge 5 is outputted, and subsequently it is changed to approach the above steady state voltage value at a time constant which is determined by the capacitor C80 and the resistors R78 and R79.
Of the two drawbacks discussed above, the drawback in a) may not be as such depending on the field of applications. However, in recently expanding applications where the sensor output signal is converted by an analog-to-digital (AD) converter into digital signal for signal processing in a microprocessor, the reference voltage of the analog-to-digital converter is often proportional to the power supply voltage, and in such a case it is required for the sensitivity of the sensor to be proportional to the power supply voltage.
As for the drawback in b) above, in applications requiring low frequency signal, the time constant of charging of the coupling capacitor is inevitably large, thus dictating a very long time until the sensor operation is stabilized after the connection of the power supply source, which is a significant problem.